Controllable mixer

ABSTRACT

A heterodyne receiver has a mixer with at least one transistor whose operating point can be varied dynamically. The quality of the output signal from the mixer is assessed in order to control the operating point. The operating point is set such that the collector current is increased when the intermodulation interference is high, thus improving the intermodulation resistance. The collector current is reduced when the intermodulation interference is low, thus reducing the transistor noise. Furthermore, the current drawn is reduced in this situation. The circuit and the method are particularly suitable for RF receivers without tunable input filters, and for receivers in which the power consumption must be low.

Nowadays, receivers for modulated radio-frequency signals are normallyin the form of heterodyne receivers. Heterodyne receivers use a mixingstage to which the input signal to be received and an oscillator signalare supplied. The oscillator signal is tunable as a function of thedesired frequency which is intended to be received. The mixing stageproduces at its output a signal which, for example, is at a lowerfrequency than the input signal. This frequency is referred to as theintermediate frequency. The input signal, after having been down-mixedto the intermediate frequency, is passed via a bandpass filter and isdemodulated in a downstream demodulator stage, in a typical receiver.The mixer is frequently preceded by a controllable amplifier whichmatches the level of the input signal to the input of the mixer. Thismeasure prevents interference signals being produced by overdriving as aresult of non-linearities in the mixing stage. On the other hand, weakinput signals are amplified to such an extent that any noise which isadded in the mixer has a negative effect on the signal-to-noise ratio.The so-called automatic gain control (AGC) thus ensures that the levelof an input signal is matched to a downstream stage.

Furthermore, the signal flowpath upstream of the mixer also frequentlycontains a tunable bandpass filter, by means of which signals which areadjacent to the useful signal are reduced or suppressed. Suppression orreduction of signals which are adjacent to the useful signal isnecessary because intermodulation interference can be caused in themixer by the proximity of the useful signal and adjacent signals.Furthermore, adjacent signals which are at a higher signal level thanthe useful signal overdrive the mixer. This is the situation when thecontrollable amplifier upstream of the mixing stage matches the level ofthe useful signal to the input of the mixer and at the same time alsoraises the adjacent signal above the maximum permissible input level ofthe mixer.

The suppression of signals which are adjacent to a useful signalinvolves a high degree of circuitry complexity. Bandpass filtersupstream of the mixer must be tunable to the tuned frequency.Furthermore, circuits or filters for suppressing adjacent signals mustbe trimmed at the factory during manufacture of receivers.

It is thus desirable to specify a circuit which has a mixer and in whichintermodulation interference is suppressed and the noise behaviour isimproved with less complex circuitry and with less need to trim circuitparts during manufacture of receivers. A further aim is to specify amethod for controlling a circuit according to the invention foroptimization of the immunity to interference.

A mixer such as this and a method such as this are specified in theindependent claims. Advantageous refinements and further developmentsare specified in the dependent claims.

The mixer according to the invention has at least one transistor whoseoperating point can be set by means of a control signal. An evaluationcircuit which is connected via a bandpass filter downstream from themixer evaluates the signal quality of the output signal. If theintermodulation interference is strong, as occurs, by way of example,when two strong signals are at closely adjacent frequencies, theoperating point of the mixer is set such that a high collector currentflows. When that collector current is high, the modulation range of thetransistor in the mixer increases. The larger modulation range isrequired for two adjacent strong signals in order to avoid the creationof intermodulation products as a result of non-linearities in thetransistor. One embodiment of the mixer makes use of an effect whichoccurs particularly in bipolar transistors. In this case, the mixinggain of the transistor at the same time decreases when high collectorcurrents occur, which are required for large input signals. When thecollector currents are low, the mixing gain rises. The collector currentis reduced for small input signals and when the level of the adjacentsignals is only low, because the requirements for the modulation rangeof the transistor are less. At the same time, a higher mixing gainoccurs in mixers designed using bipolar transistors, and this isdesirable for low input signals. Furthermore, the noise from thetransistor is also reduced when the collector currents are low.

During reception of digitally coded signals, in which error correctionis possible, the signal quality can be determined in a simple manner byevaluation of the error rate. However, other possible ways to determinethe signal quality are also feasible, depending on the type ofmodulation used and on the input signal, for example analysis of thefrequency spectrum of the output signal from the mixing stage.

The invention advantageously allows dynamic matching of the mixercharacteristic to the respective reception situation, taking intoaccount signals adjacent to the received signal. This makes it possibleto improve the resistance to interference, with less circuitrycomplexity.

A receiver circuit according to the invention for reception of digitalsignals does not require a tunable bandpass filter at the input. Thedigital circuit evaluates the error rate of the received signal, andcontrols the characteristic of the mixing stage accordingly.

In particular, the circuit and the method are also suitable for mobileappliances or other appliances in which the current drawn should beminimal (intelligent power management). Dynamic adjustment of thecharacteristic of the transistor in the mixing stage makes it possibleto reduce the collector current, and thus the entire current that isdrawn, when the reception situation is good and the signal levels arelow. The reduced current drawn advantageously makes it possible toreduce the effort for heat dissipation in the circuitry. This alsosimplifies integration of the mixer and of further components, such as ademodulator, in a single integrated circuit.

In a further development, initial values for operation of the mixer inthe receiver are stored in a memory. These initial values include, forexample, information about the modulation method, the code rate and/orthe symbol rate. The modulation method may, inter alia, comprise phasemodulation methods such as BPSK (Binary Phase Shift Keying), QPSK(Quadrature Phase Shift Keying), 8PSK (8 Phase Shift Keying), orcombined phase amplitude modulation methods such as QAM (QuadratureAmplitude Modulation), or else frequency modulation methods such as OFDM(Orthogonal Frequency Division Multiplex).

The initial values are used as the basis for assessment of the currentsignal quality, and the operating point of the mixer is set so as toachieve at least a desired minimum signal quality. Values for a desiredminimum signal quality are advantageously likewise stored in the memory,and the desired minimum signal quality may differ, depending on themodulation method that is used. In this case, values for the desiredminimum signal quality are stored for each modulation method. In thecase of digitally coded signals, the signal quality is inverselyproportional to the error rate.

In a further development, individual optimization routines are storedfor each of the different initial values mentioned above. Theoptimization routines are then used to optimize the setting of themixer.

The theoretical principle of the invention can be derived from analysisof the non-linear transmission characteristic of a four-pole network(illustrated here only up to the 3rd order):y=a·x+b·x ² +c·x ³   (1)

A two-tone signal x(t) is passed to the four-pole network with thetransfer function as in equation (1):x(t)=μ·sin( ω ₁ ·t)+ν·sin( ω ₂ ·t)   (2)

Equation (3) is obtained by substitution of (2) and (1): $\begin{matrix}{y = {{\frac{1}{2}{b \cdot u^{2}}} + {\frac{1}{2}{b \cdot v^{2}}}}} & \lbrack 1\rbrack \\{{{+ a} \cdot u \cdot {\sin\left( {\varpi_{1} \cdot t} \right)}} + {a \cdot v \cdot {\sin\left( {\varpi_{2} \cdot t} \right)}} + {\frac{3}{4}{c \cdot u^{3} \cdot {\sin\left( {\varpi_{1} \cdot t} \right)}}} + {\frac{3}{4}{c \cdot v^{3} \cdot {\sin\left( {\varpi_{2} \cdot t} \right)}}}} & \lbrack 2\rbrack \\{{{+ \frac{3}{2}}{c \cdot u \cdot v^{2} \cdot {\sin\left( {\varpi_{1} \cdot t} \right)}}} + {\frac{3}{2}{c \cdot u^{2} \cdot v \cdot {\sin\left( {\varpi_{2} \cdot t} \right)}}}} & \lbrack 3\rbrack \\{{{+ b} \cdot u \cdot v \cdot {\cos\left( {\left( {\varpi_{1} - \varpi_{2}} \right)t} \right)}} - {b \cdot u \cdot v \cdot {\cos\left( {\left( {\varpi_{1} + \varpi_{2}} \right)t} \right)}}} & \lbrack 4\rbrack \\{{{- \frac{1}{2}}{b \cdot u^{2} \cdot {\cos\left( {2{\varpi_{1} \cdot t}} \right)}}} - {\frac{1}{2}{b \cdot v^{2} \cdot {\cos\left( {2{\varpi_{2} \cdot t}} \right)}}}} & \lbrack 5\rbrack\end{matrix}$

In the equation above, the equation parts are as follows:

[1] represents the DC component,

[2] represents the linear component,

[3] represents the cross-modulation component,

[4] represents the intermodulation IM2,

[5] represents the square component (twice the frequencies of thesignals ω₁ and ω₂) ;

[6] represents the intermodulation IM3, and

[7] represents the cubic component (three times the frequencies of thesignals ω₁ and ω₂).

The factors b and c in (1) can be controlled by suitable control of themixer characteristic such that the non-linear components IM2 [4] and IM3[6] are variable. The characteristic, and hence the factors b and c, arecontrolled via the control input in the circuit according to theinvention.

The invention will be described in the following text with reference tothe drawing, in which:

FIG. 1 shows a receiver according to the prior art,

FIG. 2 shows a receiver with a mixer according to the invention,

FIG. 3 shows a first schematic illustration of a mixer according to theinvention,

FIG. 4 shows a second schematic illustration of a mixer according to theinvention,

FIG. 5 shows an illustration of the intermodulation immunity as afunction of the collector current for a transistor,

FIG. 6 shows a schematic illustration of the input and output signals ofa mixer for different transistor operating points.

Identical or similar elements are provided with the same referencesymbols in the figures.

FIG. 1 shows a schematic block diagram of a receiver according to theprior art. An input signal RF_(in) is passed to a tunable bandpassfilter 1. The tunable bandpass filter 1 is used for selection of thedesired input signal, and for suppression of possible adjacent signals.The signal is passed from the tunable bandpass filter 1 to avariable-gain amplifier 2. The amplifier 2 is connected to a mixer 3.The mixer 3 is also supplied with the signal at a variable frequencyfrom an oscillator 4. An intermediate frequency signal IF is produced atthe output of the mixer 3 at a frequency which is lower than thefrequency of the input signal RF_(in). The intermediate frequency signalIF is passed to a bandpass filter 6 at a fixed mid-frequency. Theintermediate frequency signal is passed from the bandpass filter 6 to acontrol circuit 7, and to a demodulator 8. The demodulated signal isproduced at an output 9 of the demodulator 8 for further processing. Thecontrol circuit 7 uses a control signal AGC to control the variableamplifier 2. This control loop ensures that the input signal RF_(in) isapplied to the mixer 3 at a suitable signal level. The demodulator 8demodulates the signal for further processing.

FIG. 2 shows a schematic block diagram of a receiver with a mixeraccording to the invention. An input signal RF_(in) is passed to anamplifier 2. The input signal is passed from the amplifier 2 to a mixer3. The mixer 3 is supplied with the signal at a variable frequency froman oscillator 4. Furthermore, the mixer 3 is supplied with a signalAGQC. An intermediate frequency signal IF is passed from the output ofthe mixer 3 to a bandpass filter 6 at a fixed mid-frequency. The signalis passed from the bandpass filter 6 to a demodulator 8. The demodulator8 demodulates the received signal, and produces it at an output 9. Thesignal at the output 9 is also passed to an evaluation circuit 7, whichassesses the signal quality and produces the monitoring signal AGQC as afunction of the quality of the received and demodulated signal, whichmonitoring signal is applied to the mixer 3. A memory 5 is connected tothe evaluation circuit for storage and for reading initial values, aswell as data obtained during operation.

FIG. 3 shows a first schematic circuit diagram of a mixer according tothe invention. A radio-frequency signal RF_(in) is passed via a couplingcapacitor 11 to the base connection of a transistor 12. The operatingpoint of the transistor 12 is set via a voltage divider comprising theresistors 13 and 14 at the base connection of the transistor 12. Acontrol voltage Us is applied to the base connection of the transistor12 via a resistor 16. The control voltage Us is derived from the signalAGQC, which is not illustrated in the figure. The operating point of thetransistor 12 can be varied by means of the control voltage Us. Acapacitor 18 and an inductance 19, connected in parallel, are connectedto an operating voltage UB at the collector connection of the transistor12. The parallel circuit formed by the capacitor 18 and the inductance19 forms an IF filter 17. The intermediate frequency signal IF is alsoproduced at the collector output of the transistor 12, and is emittedvia a coupling capacitor 23. An emitter resistor 22 is connected toearth at the emitter connection of the transistor 12. Furthermore, theemitter connection of the transistor 12 is supplied via a couplingcapacitor 21 with the signal LO at a variable frequency from anoscillator which is not illustrated in the figure.

FIG. 4 shows a second schematic illustration of a mixer according to theinvention. The mixer in FIG. 4 is suitable for processing balancedsignals. The negative mathematical sign in the annotation of the signalsindicates that the signals are in antiphase. A signal RF_(in) is passedvia a coupling capacitor 11 to the base connections of two transistors26 and 29. The base connections of the transistors 26 and 29 areconnected to earth via a resistor 14. An antiphase signal −RF_(in) ispassed via a coupling capacitor 111 to the base connections of twotransistors 27 and 28. The base connections of the transistors 27 and 28are connected to earth via a resistor 114. The transistor pairs 26 and27 as well as 28 and 29 are connected as differential amplifiers. Theemitters of the transistor pairs 26 and 27 as well as 28 and 29 arerespectively connected to one another. The connected emitter connectionsof the transistors 26 and 27 are connected to earth via a resistor 30and a capacitor 31. The connected emitter connections of the transistorpair 28 and 29 are likewise connected to the capacitor 31 via a resistor130, and are connected to earth via this capacitor 31. The signal LOfrom an oscillator is applied to the connected emitter connections ofthe transistors 26 and 27 via a coupling capacitor 21. The antiphasesignal −LO is applied to the connected emitter connections of thetransistor pair 28 and 29 via a coupling capacitor 121. A controlvoltage Us, which is derived from the signal AGQC (which is notillustrated in the figure), is connected between the resistor 30 and thecapacitor 31. The operating points of the differential amplifiers formedby the transistor pairs 26 and 27 as well as 28 and 29 can be adjustedby means of the control voltage U_(S). The collector connections of thetransistors 2 6 and 28 are connected to one another. The collectorconnections of the transistors 27 and 29 are likewise connected to oneanother. The intermediate frequency signal IF and the associatedantiphase signal −IF can be tapped off at the connected collectorconnections of the transistors via output capacitors 23 and 123. Aparallel circuit formed by an inductance 19 and a capacitor 1B iscoupled between the connected collector connections of the transistorpairs 26 and 28 as well as 27 and 29. The circuit formed by theinductance and the capacitor forms an intermediate frequency filter 17.In FIG. 4, the inductance 19 is formed from two series-connectedinductance elements, at whose centre connection the supply voltage forthe differential amplifiers is fed. Feeding the supply voltage via thecentre connection avoids the direct current having any influence on theinductance.

FIG. 5 shows an illustration of the intermodulation resistance IM3 of atransistor, as a function of the collector current. The family ofcharacteristics clearly shows that the intermodulation immunity level isa function of the collector current when the collector/emitter voltageis constant.

By way of example, FIG. 6 shows a simplified schematic illustration ofthe input and output signals of a mixer for different operating pointsof a transistor. FIG. 6 a shows two inputs signals RF_(use) and RF_(adj)with the same signal levels. The useful signal RF_(use) is at afrequency of 205 MHz. The adjacent signal RF_(adj) is at a frequency of214 MHz. It is assumed that the mixer oscillator is operating at afrequency of 200 MHz. The intermediate frequency signals IF_(use) andIF_(adj), on the one hand, and the undesirable intermodulation productIF_(int), on the other hand, are produced in the mixer. The usefulintermediate frequency signal IF_(use) is at a frequency of 5 MHz (205MHz−200 MHz), the adjacent intermediate frequency signal is at frequencyof 14 MHz (214 MHz−200 MHz), and the interference intermediate frequencysignal IF_(int), which is formed by intermodulation, is at a frequencyof 4 MHz (200 MHz−(2×205 MHz−214 MHz)). FIG. 6 b shows examples ofuseful, adjacent and interference intermediate frequency signals. Thediagram in FIG. 6 b is based on the assumption that the mixingtransistor is set to a high mixing gain. The three intermediatefrequency output signals are at relatively high levels, with theinterference intermediate frequency signal IF_(int) being at only aslightly lower level than the useful intermediate frequency signalIF_(use). The intermodulation separation, which is the separationbetween the useful signal and the interference signal, is, by way ofexample, assumed to be X dBc, where dBc represents a weightedmeasurement. FIG. 6 c shows the output signals from the mixer when themixing gain of the mixer transistor is lower. The useful intermediatefrequency signal IF_(use) is at a lower level than in FIG. 6 b. Theinterference intermediate frequency signal IF_(int) has not been reducedto the same extent as the useful intermediate frequency signal. Theintermodulation separation was increased considerably in comparison tothe example in FIG. 6 b, and is Y dBc. In this case, Y dBc is greaterthan X dBc.

1. RF-circuit including a controllable mixer having at least onetransistor, to which an oscillator signal and an input signal aresupplied, with the input signal comprising a useful signal and furthersignals, and with an output signal being produced as an output of themixer, wherein a controller is provided, which applies a control signalto the mixer as a function of the signal quality of the output signal,wherein the operating point of the at least one transistor can be set bymeans of the control signal, in which case the intermodulation immunityand/or the noise in the output signal can be varied as a function of theoperating point of the at least one transistor wherein a controllableportion of the overall gain of the RF-circuit is determined by theoperating point of the at least one transistor of the mixer. 2.Controllable mixer according to claim 1, wherein a demodulator which isconnected downstream from the mixer, and an evaluation circuit areprovided for assessment of the signal quality of the output signal. 3.Controllable mixer according to claim 2, wherein the evaluation circuitassesses the error rate of a digitally coded signal.
 4. Controllablemixer according to claim 1, wherein a memory is provided for recordinginitial values, on the basis of which the signal quality can be assessedand optimized.
 5. Controllable mixer according to claim 4, wherein theinitial values comprise information about a desired minimum signalquality, the symbol rate, the code rate, and/or the modulation method,and optimization routines for reception optimization can be selected asa function of the initial values.
 6. Method for controlling a mixer in areceiver having at least one transistor to which an oscillator signaland an input signal are supplied, with the input signal comprising auseful signal and further signals, and with an output signal beingproduced as an output of the mixer, the method comprising the followingsteps: assessing the signal quality of the output signal; setting theoperating point of the at least one transistor as a function of thequality of the output signal; wherein the intermodulation immunityand/or the noise of the at least one transistor are set by means of theoperating point of the at least one transistor wherein the method isfurther comprised by setting a controllable portion of the overall gainof the RF-circuit in by setting the operating point of the at least onetransistor of the mixer.
 7. Method according to claim 6, wherein theerror rate of a digitally coded signal is evaluated in order to assessthe signal quality.
 8. Method according to claim 6, wherein initialvalues which are stored at the start are selected in order to assess thesignal quality and in order to set the operating point of thetransistor.
 9. Method according to claim 8, wherein different initialvalues and/or optimization routines are selected for differentmodulation methods, code rates and/or symbol rates.